Transmission apparatus for a wireless device using delta-sigma modulation

ABSTRACT

A transmission apparatus for a wireless device, comprising: an antenna for receiving an original signal and for backscattering a modulated signal containing information from the wireless device; a variable impedance coupled to the antenna, the variable impedance having an impedance value; a delta-sigma modulator coupled to the variable impedance for modulating the impedance value, and thereby a backscattering coefficient for the antenna, in accordance with the information to generate the modulated signal; and, a decoder coupled to the delta-sigma modulator for generating the impedance value from the information.

This application is continuation of U.S. patent application Ser. No.15/513,100, filed on Mar. 21, 2017, entitled TRANSMISSION APPARATUS FORA WIRELESS DEVICE USING DELTA-SIGMA MODULATION, which is a 35 U.S.C. 371of International Application of PCT PCT/CA2014/000745, entitledTRANSMISSION APPARATUS FOR A WIRELESS DEVICE USING DELTA-SIGMAMODULATION, filed on Oct. 16, 2014, which claims priority from U.S.patent application Ser. No. 14/493,262, filed Sep. 22, 2014, andincorporated herein by reference.

FIELD OF THE INVENTION

This invention relates to the field of radio frequency identificationsystems, and more specifically, to transmission apparatus for wirelessdevices (e.g., tags) in backscattered and inductively coupled radiofrequency identification systems.

BACKGROUND OF THE INVENTION

Radio frequency identification (“RFID”) systems have become very popularin a great number of applications. A typical RFID system 100 is shown inFIG. 1. The RFID system 100 includes an application system 110, a reader120, and a tag 130. When the tag 130 appears in the operational range ofthe reader 120, it starts receiving both energy 140 and data 150 via itsantenna 133 from the reader 120 via its transmitter/receiver 121 andantenna 123. A rectify circuit 131 in the tag 130 collects and storesthe energy 140 for powering the other circuits (e.g., control/modulator132) in the tag 130. After collecting enough energy 140, the tag 130 mayoperate and send back pre-stored data to the reader 120. The reader 120then passes the received response data via a communications interface160 to the server system/database 111 of the application system 110 forsystem applications.

The tags 130 in RFID system 100 may be classified into passive andactive types according to the power provisions of the tags. Passive tagsdo not have their own power supply and therefore draw all power requiredfrom the reader 120 by electromagnetic energy received via the tag'santenna 133. In contrast, active tags incorporate a battery whichsupplies all or part of the power required for their operation.

A typical transmission method of energy 140 and data 150 between areader 120 and a tag 130 in a RFID system 100 is by way of backscattercoupling (or backscattering). The antenna 123 of the reader 120 couplesenergy 140 to the tag 130. By modulating the reflection coefficient ofthe tag's antenna 133, data 150 may be transmitted between the tag 130and the reader 120. Backscattering, as shown in FIG. 2, is typicallyused in microwave band RFID systems. Power P_(in) is emitted from thereader's antenna 123. A small proportion of P_(in) is received by thetag's antenna 133 and is rectified to charge the storing capacitor inthe tag 130 for serving as a power supply. After gathering enoughenergy, the tag 130 begins operating. A portion of the incoming powerP_(in) is reflected by the tag's antenna 133 and returned as powerP_(return). The reflection characteristics may be influenced by alteringthe load connected to the antenna 133. In order to transmit data 150from the tag 130 to the reader 120, a transistor is switched on and offin time with the transmitted data stream. The magnitude of the reflectedpower P_(return) may thus be modulated and picked up by the reader'santenna 123.

Amplitude shift keying (“ASK”) modulation is typically used in RFIDsystems 100. In ASK modulation, the amplitude of the carrier is switchedbetween two states controlled by the binary transmitting code sequence.Also, in some applications, phase shift keying (“PSK”) modulation isalso used. However, arbitrary complex type modulations are generally notused in current RFID backscattering systems. Here complex typemodulations are ones that are normally expressed as I+jQ, where I is thein-phase component, Q is the quadrature component, and j is the squareroot of −1.

For reference, the beginnings of RFID use may be found as far back asWorld War II. See for example, Stockman H., “Communication By Means ofReflected Power,” Proc. IRE, pp. 1196-1204, October 1948. Passive andsemi-passive RFID tags were used to communicate with the reader by radiofrequency (“RF”) backscattering. In backscattering RFID systems, anumber of tags 130 interact with a main reader device 120 as shown inFIG. 3. The reader 130 is used to: (i) power up the tags 130 via thepower of the RF signal; (ii) transfer data to the tags 130; and, (iii)read information from the tags 130.

Typically, a link budget exists between the reader 120 and the tag 130.The tag 130 communicates with the reader 120 by backscattering the RFsignal back to the reader 120 using either ASK or PSK modulation. Oneadvantage of the backscattering method is that it does not need togenerate an RF carrier on chip within the tag 130, thus it requires lesspower, less complexity, and less cost. A typical block diagram of abackscattering transmission apparatus 400 for a tag 130 is shown in FIG.4. In FIG. 4, Z_(ant) is the impedance of the antenna 133 and Z_(o) is afixed impedance which is in parallel with a switch 410. The reflectioncoefficient Γ is given by the equation:

$\Gamma = \frac{Z_{0} - Z_{ant}}{Z_{0} + Z_{ant}}$

With the switch 410 on (i.e., closed), Γ=1. When the switch is off(i.e., open), Γ=0. By turning the switch 410 on and off, an ASK signal420 is generated as shown in FIG. 4.

PSK signals may also be generated using a similar set up. This is shownin the transmission apparatus 500 illustrated in FIG. 5. Here, thereflection coefficient Γ is given by the equation:

$\Gamma = \frac{\left( {Z_{i} - Z_{0}} \right) - Z_{ant}}{\left( {Z_{i} - Z_{0}} \right) + Z_{ant}}$

Here, Z_(i) is an impedance that is switched in as per FIG. 5. So,depending on the position of the switch 410, 510, backscattering isdesigned to produce either an ASK signal 420 or a PSK signal 520.

As shown in FIG. 6, using backscattering techniques, each tag 130 sendsRF signals 610 on the same carrier 620 and hence overlapping the RFspectrum of other tags 130. This poses a challenge which respect toavoiding data collisions between all of the tags 130. In currentsystems, these collision issues are solved via the communicationprotocol used between the reader 120 and the tags 130.

In Thomas S., Reynolds S. Matthew, “QAM Backscatter for Passive UHF RFIDTags”, IEEE RFID, p. 210, 2010 (Thomas et al.), the generation of fourquadrature amplitude modulation (“QAM”) signals was proposed in which anumber of Γ values are switched in and out.

There are several problems with prior tag transmission apparatus. Forexample, systems such as that proposed by Thomas et al. are limited inthe nature of signals that they can backscatter. That is, any arbitrarysignal cannot be transmitted. For example, if the QAM signal is firstfiltered via a filter, Thomas et al.'s system cannot transmit a filteredversion of the QAM signal. As another example, if the signal is simply asine wave or a Gaussian minimum shift keying (“GMSK”) signal, Thomas etal.'s system cannot be used to transmit this signal. As a furtherexample, Thomas et. al.'s system cannot transmit single side bandsignals.

A need therefore exists for an improved transmission apparatus forwireless devices (e.g., tags) in backscattered and inductively coupledradio frequency identification systems. Accordingly, a solution thataddresses, at least in part, the above and other shortcomings isdesired.

SUMMARY OF THE INVENTION

According to one aspect of the invention, there is provided atransmission apparatus for a wireless device, comprising: an antenna forreceiving an original signal and for backscattering a modulated signalcontaining information from the wireless device; a variable impedancecoupled to the antenna, the variable impedance having an impedancevalue; a delta-sigma modulator coupled to the variable impedance formodulating the impedance value, and thereby a backscattering coefficientfor the antenna, in accordance with the information to generate themodulated signal; and, a decoder coupled to the delta-sigma modulatorfor generating the impedance value from the information.

BRIEF DESCRIPTION OF THE DRAWINGS

Features and advantages of the embodiments of the present invention willbecome apparent from the following detailed description, taken incombination with the appended drawings, in which:

FIG. 1 is a block diagram illustrating a radio frequency identification(RFID) system in accordance with the prior art;

FIG. 2 is a block diagram illustrating transmission of energy and databetween a reader and a tag in a RFID system in accordance with the priorart;

FIG. 3 is a block diagram illustrating communications between a readerand multiple tags in an RFID system in accordance with the prior art;

FIG. 4 is a block diagram illustrating a transmission apparatus for atag for backscattering ASK and/or on-off keying (“OOK”) signals inaccordance with the prior art;

FIG. 5 is a block diagram illustrating a transmission apparatus for atag for backscattering PSK signals in accordance with the prior art;

FIG. 6 is a block diagram illustrating multiple tags communicating backto a reader using the same frequency spectrum in accordance with theprior art;

FIG. 7A is a block diagram illustrating a transmission apparatus for awireless device for backscattering signals to a reader based on adigital wave form input in accordance with an embodiment of theinvention;

FIG. 7B is a block diagram illustrating a variable impedance circuit forthe transmission apparatus of FIG. 7A in accordance with an embodimentof the invention;

FIG. 8 is a graph illustrating the relationship between Gamma (Γ) andZ_(i) in accordance with an embodiment of the invention;

FIG. 9 is a block diagram illustrating a transmission apparatus with anadder for a wireless device for backscattering arbitrary modulatedsignals to a reader based on I and Q data input in accordance with anembodiment of the invention;

FIG. 10A is a block diagram illustrating inductive coupling between areader and a wireless device in a RFID system in accordance with anembodiment of the invention;

FIG. 10B is a block diagram illustrating an equivalent circuit for theRFID system of FIG. 10A in accordance with an embodiment of theinvention; and,

FIG. 11 is a block diagram illustrating a transmission apparatus usinginductive coupling for a wireless device for transmitting signals to areader based on a digital waveform input in accordance with anembodiment of the invention.

It will be noted that throughout the appended drawings, like featuresare identified by like reference numerals.

DETAILED DESCRIPTION OF THE EMBODIMENTS

In the following description, details are set forth to provide anunderstanding of the invention. In some instances, certain software,circuits, structures and methods have not been described or shown indetail in order not to obscure the invention. The term “apparatus” isused herein to refer to any machine for processing data, including thesystems, devices, and network arrangements described herein. The term“wireless device” is used herein to refer to RFID tags, RFIDtransponders, cellular telephones, smart phones, portable computers,notebook computers, or similar devices. The present invention may beimplemented in any computer programming language provided that theoperating system of the data processing system provides the facilitiesthat may support the requirements of the present invention. Anylimitations presented would be a result of a particular type ofoperating system or computer programming language and would not be alimitation of the present invention. The present invention may also beimplemented in hardware or in a combination of hardware and software.

FIG. 7A is a block diagram illustrating a transmission apparatus 800 fora wireless device 130 for backscattering signals to a reader 120 basedon a digital wave form input 830 in accordance with an embodiment of theinvention. And, FIG. 7B is a block diagram illustrating a variableimpedance 810 circuit for the transmission apparatus 800 of FIG. 7A inaccordance with an embodiment of the invention. The present inventionprovides a method and apparatus for generating complex waveforms forpassive and semi-passive RFID systems 100. The complex wave forms maygenerate any type of complex modulation signals such as 8-constellationphase shift keying (“8 PSK), orthogonal frequency-division multiplexing(“OFDM”), or n-constellation quadrature amplitude modulation (“nQAM”).The method and apparatus may also be used to generate frequency channelsfor each wireless device 130. According to one embodiment, thetransmission apparatus (e.g., 800) includes an antenna 133 coupled to avariable impedance 810 having an array of impedances (e.g., the firstimpedance Z₁ and the second impedance Z₂ in FIG. 7B) that are switchedon or off (via the first switch S₁ and the second switch S₂ in FIG. 7B,respectively) via a backscattering decoder 820 and a delta-sigma (ΔΣ)modulator 840 in the wireless device 130. The signal 830 applied to theinput of the decoder 820 may consist of any type of digital signal. Thetransmission apparatus 800 may include a processor 880 for controllingthe decoder 820, delta-sigma (ΔΣ) modulator 840, and variable impedance810, memory 890 for storing information (e.g., digital waveforms 830),and related hardware and software as is known to one of skill in theart.

FIG. 8 is a graph illustrating the relationship between Gamma (Γ) andZ_(i) in accordance with an embodiment of the invention. Here, Γ is thereflection coefficient and Z_(i) is the impedance seen by the antenna133. The reflection coefficient is directly proportional to the digitalwave form 830. According to one embodiment of the invention, forbackscattering RF applications, the reflection or backscatteringcoefficient Gamma (Γ) is given by:Γ=αe ^(jϕ) ^(i)where φ_(i) is the phase, a is the magnitude of the reflectioncoefficient, and j is the square root of −1. The back scatteringimpedance (i.e., the impedance seen by the antenna 133) is then givenby:

$Z_{i} = \frac{Z_{0}\left( {1 + {\alpha\; e^{j\;\phi_{i}}}} \right)}{\left( {1 - {\alpha\; e^{j\;\phi_{i}}}} \right)}$where Z_(o) is a constant (typically 50 ohms) and Z_(i) is the backscattering impedance value.

Assuming the phase is zero:

$Z_{i} = \frac{Z_{0}\left( {1 + \alpha} \right)}{\left( {1 - \alpha} \right)}$

If s(t) is a signal (e.g., a sine wave) that is to be sent to the reader120, it must be directly related to a(t) (e.g., s(t) is directlyproportional to a(t)) and thus Γ. This produces an impedance value Z_(i)that varies with time.

In this embodiment, the signal s(t) would be backscattered back to thereader 120 by the wireless device 130. In the transmission apparatus 800shown in FIG. 7A, N-bits 821 are applied to the variable impedance 810via the delta-sigma (ΔΣ) modulator 840 such that the impedance valueZ_(i) is encoded as shown in FIG. 8. Here, the variable impedance 810has N states. If there are any errors in the encoding or imperfectionsin encoding of Z_(i), these may be corrected within the reader 120. Thisis possible if for some time the signal s(t) is known by the reader 120.The reader 120 than may add distortion to the incoming signal to correctfor these imperfections.

Referring again to FIGS. 7A and 7B, the digital wave form information830 (e.g., N-bit information) is applied to an element such as a decoder820 that converts a scatter value or reflection coefficient Γ for theinformation 830 into an impedance value Z_(i). This impedance valueZ_(i) (e.g., N-bits 821) is then applied to a delta-sigma (ΔΣ) modulator840 which controls the switches S₁, S₂ in the variable impedance 810 toswitch between respective impedances Z₁, Z₂ based on the output of thedelta-sigma (ΔΣ) modulator 840. For example, if the output of thedelta-sigma (ΔΣ) modulator 840 is “1”, the impedance value Z_(i) is setto the value of the second impedance Z₂. If the output of thedelta-sigma (ΔΣ) modulator 840 is “0”, the impedance value Z_(i) is setto the value of the first impedance Z₁. For example, impedance values of116 ohms for the second impedance Z₂ and 21 ohms for the first impedanceZ₁ may correspond to reflection coefficients Γ of 0.4 and −0.4,respectively. If the desired reflection coefficient Γ is zero, thedecoder 820 may determine an impedance value Z_(i) of 50 ohms (as perthe graph shown in FIG. 8). The delta-sigma (ΔΣ) modulator 840 nowgenerates an output that produces an average impedance value for thevariable impedance 810 of 50 ohms by switching between the 21 and 116ohms impedances Z₂, Z₁.

As shown in FIG. 7B, the variable impedance 810 circuit may be made upof an array of impedances Z₁, Z₂ that are switched in and out byrespective switches S₁, S₂ depending on the digital decoder 820 anddelta-sigma (ΔΣ) modulator 840. Also, the variable impedance 810 may becontrolled via an analog signal, that is, after the Gamma to Z_(i)decoder 820 and delta-sigma (ΔΣ) modulator 840, a digital to analogconverter (“DAC”) (not shown) may be added to drive the variableimpedance 810.

The delta-sigma (ΔΣ) modulator 840 may be of variable design. Forexample, according to one embodiment, the delta-sigma (ΔΣ) modulator 840may include or be a low-pass delta-sigma (ΔΣ) modulator. According toanother embodiment, the delta-sigma (ΔΣ) modulator 840 may include or bea band-pass delta-sigma (ΔΣ) modulator. According to one embodiment, thedelta-sigma (ΔΣ) modulator 840 may be a single bit delta-sigma (ΔΣ)modulator.

The delta-sigma (ΔΣ) modulator 840 generates an output bit stream thatrepresents the input data 821 from a DC level to some predetermineddesign bandwidth. Beyond the predetermined design bandwidth, quantizednoise of the delta-sigma (ΔΣ) modulator 840 may increase until, at somedesign cutoff point, the signal may be deemed to have too muchquantization noise. According to one embodiment, one or more filters maybe included in the variable impedance 810 circuit to filter out-of-bandnoise output from the delta-sigma (ΔΣ) modulator 840. The variableimpedance 810 circuit has an output electrically connected to theantenna 133. The delta-sigma (ΔΣ) modulator 840 is coupled to an inputto the variable impedance 810 circuit to digitally control the output ofthe variable impedance 810 circuit such that the reflection coefficientΓ of the antenna 133 may be adjusted by changing the impedance valueZ_(i) of the variable impedance 810 circuit. According to oneembodiment, the output of the delta-sigma (ΔΣ) modulator 840 switchesthe impedance value Z_(i) of the variable impedance 810 between at leasttwo states or impedance values Z_(i).

According to one embodiment, the delta-sigma (ΔΣ) modulator 840 may beof any order based on the bandwidth of the signals being applied to it.In addition, the clock applied to the delta-sigma (ΔΣ) modulator 840 mayset the over-sampling rate.

FIG. 9 is a block diagram illustrating a transmission apparatus 1000with an adder 1050 for a wireless device 130 for backscatteringarbitrary modulated signals to a reader 120 based on I and Q data input1030 in accordance with an embodiment of the invention. According to oneembodiment, the digital waveform 830 may be in-phase (“I”) andquadrature (“Q”) data 1030 as shown in FIG. 9. In FIG. 9, a digitalsignal generator (“DSS”) 1040 may optionally up-convert (or offset) theI and Q data 1030. For example, the DSS 1040 may provide sine (orcosine) and cosine (or sine) signals 1070 that are applied to I and Qdata by respective mixers 1071. Alternatively, the DSS 1040 may generatea constant value that is multiplied onto the I and Q data (i.e., themixers 1071 act as gain elements). The Gamma to Z_(i) decoder 1020receives the up-converted (or offset) I and Q data and applies it to thevariable impedance 1010 via the delta-sigma (ΔΣ) modulator 1080. Thevariable impedance 1010 may be made up of an array of impedances thatare switched in or out (e.g., a parallel array of impedances withrespective switches).

Summarizing the above, and referring again to FIGS. 7A and 7B, accordingto one embodiment an antenna 133 is used to backscatter an incomingradio frequency signal coming from a reader 120. The antenna 133 iselectrically coupled to an array of impedance devices Z₁, Z₂ connectedto switches S₁, S₂. The array of impedance devices (e.g., variableimpedance 810) may be digitally controlled by a digital block (e.g.,decoder 820) and a delta-sigma (ΔΣ) modulator 840 that are driven by anarbitrary N-bit digital waveform (e.g., 830). The digital block 820presents an output to the array of impedances 810 via the delta-sigma(ΔΣ) modulator 840 that is related to the N-bit digital waveform 830. Achange in the impedance value of the array of impedances 810backscatters the incoming radio frequency signal thus producing a directup-converted version of the output of the digital waveform 830 withrespect to the incoming radio frequency. The output of the digital block820 and delta-sigma (ΔΣ) modulator 840 switches the array of impedances810 between various states, which changes the characteristics of thereflection coefficient F. The signal 830 applied to the digital block820 may take the form of any complex modulation signal, for example,GMSK, nPSK, 8 PSK, nQAM, OFDM, etc., and such signals may be offset fromthe incoming radio frequency signal by a frequency +/−ω.

Referring again to FIG. 9, the input 1030 to the digital block 1020 mayalternate between in-phase (i.e., I) and quadrature (i.e., Q) signalsvia a control signal, for example. Also, the array of impedances 1010may switch between backscattering coefficients that are 90 degreesoffset from each other depending on whether the data is I or Q data. Forexample, if the I signals would produce backscattering coefficients attheta degrees then the Q signals would produce backscatteringcoefficients that are theta+90 degrees. The control signal may be aclock signal. The signals 1070 applied to the I and Q signals 1030 bythe DSS 1040 may take the form of a direct current (“DC”) signal (i.e.,no frequency offset) or of sine and cosine waves at a selected frequency(i.e., to give a frequency offset of ω). The I and Q signals applied tothe digital block 1020 may be adjusted to compensate for any errors inthe impedance array 1010, the delta-sigma (ΔΣ) modulator 1080, or thedigital block 1020. The array of impedances 1010 may include somefiltering characteristics to filter off some of the digital block's 1020or delta-sigma (ΔΣ) modulator's 1080 out-of-band noise. And, the reader120 used to detect the backscattered signal from the wireless device 130may compensate for any errors generated within the impedance array 1010,the digital block 1020, or the delta-sigma (ΔΣ) modulator 1080.

FIG. 10A is a block diagram illustrating inductive coupling between areader 120 and a wireless device 130 in a RFID system 1300 in accordancewith an embodiment of the invention. FIG. 10B is a block diagramillustrating an equivalent circuit 1310 for the RFID system 1300 of FIG.10A in accordance with an embodiment of the invention. And, FIG. 11 is ablock diagram illustrating a transmission apparatus 1400 using inductivecoupling for a wireless device 130 for transmitting signals to a reader120 based on a digital waveform input 1430 in accordance with anembodiment of the invention.

According to one embodiment, communication between the reader 120 andthe wireless device 130 may occur by sensing inductive loading changesin the reader 120. Here, the reader 120 communicates with the wirelessdevice 120 via magnetic or inductive coupling. This is shown in FIGS.10A and 10B. FIGS. 10A and 10B show the basic principle of an inductivecoupled RFID system 1300. For inductive coupled systems 1300, theunderlying coils are defined by their size. It is known that a couplingsystem of two coils 1320, 1330 may be represented by an equivalenttransformer. The connection between these two coils 1320, 1330 is givenby the magnetic field (B) and the underlying value to describe thisconnection is the mutual inductance (M) and/or the coupling factor (k).

The law of Biot and Savart is given by:

$\overset{\rightharpoonup}{B} = {\frac{\mu_{0}i_{1}}{4\pi}{\int_{S}\frac{\overset{\rightarrow}{ds}x\overset{\rightarrow}{x}}{\left| \overset{\rightarrow}{x} \right|^{3}}}}$

This allows the calculation of the magnetic field at every point as afunction of the current, i₁, as well as the geometry. Here, μ_(o) is thepermeability, x is the distance, and S describes the integration-pathalong the coil. Furthermore, the mutual inductance and the couplingfactor are given by:

$M = {\int_{A_{2}}{\frac{B\left( i_{1} \right)}{i_{1}}{dA}_{2}}}$$k = \frac{M}{\sqrt{L_{1}L_{2}}}$

In these equations, A₂ describes the area of the second coil and L₁ andL₂ are the inductances of the two coils 1320, 1330. The distance xbetween the reader-coil 1320 and transponder-coil 1330 also determinesthe coupling factor. The equivalent model for this coupling is shown inFIG. 10B. The impedance value Z_(i) as seen by the reader 120 isdirectly related to the admittances Y1 and Y2. The admittances Y1 and Y2are either modulated via amplitude (e.g., ASK) or in phase (e.g., PSK).The admittances Y1 and Y2 may also be modulated using multi-phase PSKand multi-amplitude ASK.

General speaking, the signal received back by the reader 120 is afunction of the impedance value changing in the wireless device 130.Once this impedance value changes, the signal seen by the reader 120 ismodified and the reader 120 can detect this.

As in the case of backscattering, as shown in FIG. 11, a variableimpedance 1410 may be modified by a decoder 1420 via a delta-sigma (ΔΣ)modulator 1440. Here, L 1405 is the inductance on the wireless deviceside. As in the case of backscattering, the same methods described abovemay be used, for example, for: (i) generating I and Q signals; (ii)general mapping from decoding to what the reader sees; and, (iii) if asignal is known by the reader, pre-distorting the signal to produce acorrected signal.

Summarizing the above, and referring again to FIG. 11, according to oneembodiment there is provided a transmission apparatus 1400 for modifyingan incoming radio frequency (RF) signal comprising: an inductive element1405; an array of impedances 1410 controlled by switches and circuitshaving an output electrically coupled to the inductive element 1405;and, at least one digital block 1420 coupled to the array of impedances1410 via a delta-sigma (ΔΣ) modulator 1440 for digitally controlling theimpedance value Z_(i) of the array of impedances 1410; wherein theincoming RF signal is modified as the coupled array of impedances 1410of the inductive element 1405 is adjusted.

The output of the decoder 1420 and delta-sigma (ΔΣ) modulator 1440 mayswitch the array of impedances 1410 between various states whichmodifies the incoming RF signal. The signal 1430 applied to the digitalblock 1420 may take the form of any complex modulation signal, forexample, GMSK, nPSK, 8 PSK, nQAM, OFDM, etc., and such signals may beoffset from the incoming radio frequency signal by a frequency +/−ω.

The input 1430 to the digital block 1420 may alternate between thein-phase (i.e., I) and quadrature (i.e., Q) signals via a controlsignal, for example. Also, the array of impedances 1410 may modify theincoming RF signal from 0 to 90 degrees offset depending on whether thedata is I or Q data. For example, if the I signal would produce animpedance value at theta degrees then the Q signal would produce animpedance value that is theta+90 degrees. The control signal may be aclock signal. The signals (e.g., 1070) applied to the I and Q signalsmay take the form of a DC signal or of sine and cosine waves at aselected frequency. The I and Q signals applied to the digital block1420 may be adjusted to compensate for any errors in the impedance array1410 due to variations in the impedance value in the array. The array ofimpedances 1410 may have some filtering characteristics to filter offsome of the DAC quantized out-of-band noise. And, the reader 120 used todetect the modulated signal may compensate for any errors generatedwithin the impedance array 1410, the digital block 1420, or thedelta-sigma (ΔΣ) modulator 1440.

Thus, according to one embodiment, there is provided a transmissionapparatus 800 for a wireless device 130, comprising: an antenna 133 forreceiving an original signal and for backscattering a modulated signalcontaining information 830 from the wireless device 120; a variableimpedance 810 coupled to the antenna 133, the variable impedance 810having an impedance value Z_(i); a delta-sigma (ΔΣ) modulator 840coupled to the variable impedance 810 for modulating the impedance valueZ_(i), and thereby a backscattering coefficient Γ for the antenna 133,in accordance with the information 830 to generate the modulated signal(e.g., an arbitrary modulated signal); and, a decoder 820 coupled to thedelta-sigma modulator 840 for generating the impedance value Z_(i) fromthe information 830.

In the above transmission apparatus 800, the variable impedance 810 maybe coupled in series with the antenna 133. The wireless device 130 maybe powered by energy 140 from the original signal. The variableimpedance 810 may include an array of impedances and respectiveswitches. The decoder 820 may include a backscattering coefficient Γ toimpedance value Z_(i) decoder. The information 830 may be an N-bitdigital waveform 830. The N-bit digital waveform 830 may be applied tothe decoder 820 and then to a delta-sigma (ΔΣ) modulator 840 to producea control signal 821 for the variable impedance 810 that is related tothe N-bit digital waveform 830. A change in the impedance value Z_(i)may backscatter the original signal to produce the modulated signal, themodulated signal being a frequency offset (e.g., up-converted) form ofthe N-bit digital waveform 830. The control signal 821 for the variableimpedance 810 may switch an array of impedances within the variableimpedance 810 which may change characteristics of the backscatteringcoefficient Γ of the antenna 133. The information 830 may be a complexmodulation signal 1030. The complex modulation signal 1030 may be offsetin frequency from the original signal. The complex modulation signal1030 may be one of a GMSK signal, a nPSK signal, a 8 PSK signal, a nQAMsignal, and an OFDM signal. The complex modulation signal 1030 may berepresented by I+jQ, where I is an inphase component, Q is a quadraturecomponent, and j is a square root of −1. The complex modulation signal1030 may alternate between an in-phase signal (I) and a quadraturesignal (Q) via a control signal. The variable impedance 810, 1010 mayswitch between backscattering coefficients that are 90 degrees offsetfrom each other depending on whether the complex modulation signal 1030is the in-phase signal (I) or the quadrature signal (Q). The controlsignal may be a clock signal. The transmission apparatus 800, 1000 mayfurther include a digital signal generator 1040. The digital signalgenerator 1040 may apply a constant value signal to the in-phase signal(I) and the quadrature signal (Q). The digital signal generator 1040 mayapply sine and cosine wave signals 1070 to the in-phase signal (I) andthe quadrature signal (Q), respectively. The complex modulation signal1030 may be a sum of an in-phase signal (I) and a quadrature signal (Q).The transmission apparatus 800, 1000 may further include a digitalsignal generator 1040. The digital signal generator 1040 may apply aconstant value signal to the in-phase signal (I) and the quadraturesignal (Q). The digital signal generator 1040 may apply sine and cosinewave signals 1070 to the in-phase signal (I) and the quadrature signal(Q), respectively. The N-bit digital waveform 830 may be adjusted tocompensate for errors in at least one of the decoder 820, thedelta-sigma (ΔΣ) modulator 840, and the variable impedance 810. Thevariable impedance 810 may include a filter for filtering noisegenerated by at least one of the decoder 820 and the delta-sigma (ΔΣ)modulator 840. The modulated signal may be an arbitrary signal. Thewireless device 120 may be a RFID tag. The original signal may bereceived from a RFID reader 120. The RFID reader 120 may be configuredto correct for errors in at least one of the decoder 820, thedelta-sigma (ΔΣ) modulator 840, and the variable impedance 810. Thetransmission apparatus 800 may further include a processor forcontrolling the transmission apparatus 800 and memory for storing theinformation 830. The delta-sigma (ΔΣ) modulator 840 may be one of alow-pass delta-sigma modulator and a band-pass delta-sigma modulator.The delta-sigma (ΔΣ) modulator 840 may be a single bit delta-sigmamodulator. And, the delta-sigma (ΔΣ) modulator 840 may switch (S₁, S₂)the impedance value Z_(i) between at least two states (Z₁, Z₂).

The above embodiments may contribute to an improved method and apparatusfor communications between wireless device 130 and reader 120 inbackscattered and inductively coupled radio frequency identificationsystems and may provide one or more advantages. For example, thewireless devices 130 of the present invention are not limited in thenature of signals that they may backscatter or inductively couple to thereader 120. In addition, the wireless devices 130 of the presentinvention allow for filtering of these signals. In addition, thedelta-sigma (ΔΣ) modulator 840 reduces the number of impedances thatneed to switch states in order to produce a signal. Furthermore, thedelta-sigma (ΔΣ) modulator 840 enables high levels of modulation with asfew as only one impedance.

The embodiments of the invention described above are intended to beexemplary only. Those skilled in this art will understand that variousmodifications of detail may be made to these embodiments, all of whichcome within the scope of the invention.

The invention claimed is:
 1. A method comprising: receiving, via anantenna of a wireless device, an original signal transmitted by adetecting device; receiving, at a decoder of the wireless device, anN-bit digital waveform representing information within the wirelessdevice, wherein the decoder is coupled to a variable impedance circuitof the wireless device; and responsive to the received N-bit digitalwaveform, modulating a variable impedance value of the variableimpedance circuit, wherein the N-bit digital waveform has a modulationtype, wherein a backscattering coefficient of the antenna changesresponsively to the modulated variable impedance value to backscatterthe original signal as a modulated signal that is a frequency offsetform of the N-bit digital waveform, and wherein the N-bit digitalwaveform received by the decoder produces a control signal for thevariable impedance circuit that is related to the N-bit digitalwaveform.
 2. The method of claim 1, wherein the variable impedancecircuit is coupled in series with the antenna.
 3. The method of claim 1,further comprising powering the wireless device with energy receivedfrom the original signal.
 4. The method of claim 1, wherein the variableimpedance circuit includes an array of impedances and respectiveswitches.
 5. The method of claim 1, wherein the decoder includes abackscattering coefficient to impedance value decoder.
 6. The method ofclaim 1, further comprising switching, via the control signal for thevariable impedance circuit, an array of impedances within the variableimpedance circuit and thereby change the modulated variable impedancevalue and characteristics of the backscattering coefficient of theantenna.
 7. The method of claim 1, further comprising adjusting theN-bit digital waveform to compensate for errors in at least one of thedecoder and the variable impedance.
 8. The method of claim 1, furthercomprising filtering, via a filter in the variable impedance circuit,noise generated by the decoder.
 9. The method of claim 1, wherein themodulated signal is an arbitrary signal.
 10. The method of claim 1,wherein the wireless device is a radio frequency identification (RFID)tag.
 11. The method of claim 1, wherein the original signal is receivedfrom an RFID reader.
 12. The method of claim 1, further comprisingcorrecting for errors in at least one of the decoder and the variableimpedance circuit.
 13. The method of claim 1, further comprising:performing the method of claim 1 with a processor; and storing the N-bitdigital waveform in a memory device.
 14. A method comprising: receiving,via an inductor coupled to an antenna, an original signal transmitted bya detecting device; transmitting by mutual inductance a modulated signalcontaining information from a wireless device; coupling a variableimpedance circuit to the inductor, the variable impedance circuit havingan impedance element with an impedance value; modulating the impedancevalue and thereby a value of the mutual inductance, comprising:controlling a decoder to generate a control signal in accordance withthe information; and coupling the decoder and thereby the control signalto the variable impedance circuit; and generating the information as acomplex modulate signal using an in-phase signal and a quadraturesignal.
 15. The method of claim 14, wherein the complex modulationsignal alternates between an in-phase signal and a quadrature signal viaa control signal.
 16. The method of claim 15, wherein the complexmodulation signal is offset in frequency from the original signal. 17.The method of claim 15, wherein the complex modulation signal is one ofa GMSK signal, a nPSK signal, a 8PSK signal, a nQAM signal, and an OFDMsignal.
 18. The method of claim 15, wherein the variable impedanceelement switches between impedance values that are 90 degrees offsetfrom each other depending on whether the complex modulation signal isthe in-phase signal or the quadrature signal.
 19. The method of claim15, further comprising: operating a digital signal generator to apply anup-convert signal to an in-phase circuit and to a quadrature circuit;operating the in-phase circuit to multiply an in-phase input signal bythe up-convert signal to generate the in-phase signal with an offset incarrier frequency; and operating the quadrature circuit to multiply aquadrature input signal by the up-convert signal to generate thequadrature signal with an offset in carrier frequency.
 20. The method ofclaim 15, further comprising: operating a digital signal generator toapply a constant value signal to an in-phase circuit and to a quadraturecircuit; operating the in-phase circuit to multiply an in-phase inputsignal by the constant value signal to generate the in-phase signal; andoperating the quadrature circuit to multiply a quadrature input signalby the constant value signal to generate the quadrature signal.
 21. Themethod of claim 15, further comprising: operating a digital signalgenerator to apply a sine wave to an in-phase circuit and a cosine waveto a quadrature circuit; operating the in-phase circuit to multiply anin-phase input signal by the sine wave to generate the in-phase signal;and operating the quadrature circuit to multiply a quadrature inputsignal by the cosine wave to generate the quadrature signal.
 22. Themethod of claim 14, wherein the complex modulation signal is a sum of anin-phase signal and a quadrature signal.
 23. The method of claim 22,further comprising: operating a digital signal generator to apply anup-convert signal to an in-phase circuit and to a quadrature circuit;operating the in-phase circuit to multiply an in-phase input signal bythe up-convert signal to generate the in-phase signal with an offset incarrier frequency; and operating the quadrature circuit to multiply aquadrature input signal by the up-convert signal to generate thequadrature signal with an offset in carrier frequency.
 24. The method ofclaim 22, further comprising: operating a digital signal generator toapply a constant value signal to an in-phase circuit and to a quadraturecircuit; operating the in-phase circuit to multiply an in-phase inputsignal by the constant value signal to generate the in-phase signal; andoperating the quadrature circuit to multiply a quadrature input signalby the constant value signal to generate the quadrature signal.
 25. Themethod of claim 22, further comprising: operating a digital signalgenerator to apply a sine wave to an in-phase circuit and a cosine waveto a quadrature circuit; operating the in-phase circuit to multiply anin-phase input signal by the sine wave to generate the in-phase signal;and operating the quadrature circuit to multiply a quadrature inputsignal by the cosine wave to generate the quadrature signal.